97 Original scientific paper  MIDEM Society Lossy and Lossless Inductance Simulators and Universal Filters Employing a New Versatile Active Block Mohammad Faseehuddin 1 , Jahariah Sampe 2 , Sadia Shireen 3 , Sawal Hamid Md Ali 4 1 Institute of Microengineering and Nanoelectronics (IMEN),UniversityKebangsaan Malaysia (UKM), Bangi, Selangor, Malaysia 2 Institute of Microengineering and Nanoelectronics (IMEN), University Kebangsaan Malaysia (UKM), Bangi, Selangor, Malaysia 3 Department of Electronics and Communication, Indus Institute of Technology and Management, Kanpur, India 4 Department of Electrical, Electronic and Systems Engineering, Universiti Kebangsaan Malaysia, Bangi, Selangor, Malaysia Abstract: In this paper six new implementations for realizing lossy and lossless active inductors are presented. The new topologies are obtained using a newly formulated active block namely, the extra X current conveyor transconductance amplifier (EXCCTA). The designed grounded inductor simulators (GIS) require only three grounded passive elements for implementation which is desirable for integrated circuit fabrication. Moreover, the inductance simulators are electronically tunable and free from matching requirements. Additionally, two structures of universal filters are also proposed. The voltage mode (VM) filter is multi input single output (MISO) structure and uses canonical number of passive components. The current mode (CM) filter is a single input multi output (SIMO) structure with grounded passive elements. The filter structures can realize all five standard filter responses without any matching conditions and have features of low/ high output impedance, low active and passive sensitivities, tunability, independent control of pole frequency and quality factor. The effect of non-idealities on the inductor and filter topologies are also analyzed. The simulations are performed in 0.18μm parameters from TSMC in Spice to validate the theoretical predictions. Experimental results using AD844 and LM13700 integrated circuits (ICs) for the proposed inductor and filter are also provided to further ascertain the feasibility of the proposed solutions. Keywords: active inductor; current mode; current conveyor; filter; tunability Izgubni in brezizgubni induktivni simulator ter univerzalen filter z z novim prilagodljivim aktivnim blokom Izvleček: Članek predstavlja šest primerov realizacije izgubnih in brez izgubnih aktivnih dušilk. Nove topologije uporabljajo nov aktivni blok; ekstra X tokovni transkonduktančni ojačevalnik (EXCCTA). Načrtani ozemljeni induktivni simulatorji (GIS) zahtevajo le tri ozemljene pasivne elemente in so elektronsko nastavljivi ter neodvisni od zahtev ujemanja. Dodatno sta predlagani dve strukturi univerzalnega filtra. Več vhodni in eno izhodni filter v napetostnem načinu uporablja standardno število pasivnih elementov. Tokovni filter je eno vhodni in več izhodni z ozemljenimi elementi. Filtri lahko realizirajo vseh pet standardnih odzivov brez prilagajanj. Imajo nizko/visoko izhodno impedanco, nizko aktivno in pasivno občutljivost, nastavljivost, neodvisno kontrolo frekvence in faktorja kvalitete. Simulacije so izvedene v 0.18 μm tehnologiji iz TSMC in SPICE za validacijo teoretičnih predvidevanj. Eksperimentalni rezultati so pridobljeni s uporabo AD844 in LM13700 integriranih vezij. Ključne besede: aktivna dušilka; tokovni način; tokovni ojačevalnik; filter; nastavljivost * Corresponding Author’s e-mail: faseehuddin03@siswa.ukm.edu.my Journal of Microelectronics, Electronic Components and Materials Vol. 48, No. 2(2018), 97 – 113 98 M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 1 Introduction The physical spiral inductors suffer from low realiz- able inductance value of the order of 1nH [1, 2], low quality factor, higher chip area requirement, excessive losses and susceptibility to process variations [1]. These drawbacks make them unsuitable for integrated circuit implementation. Since inductors are inevitable com- ponents given their broad spectrum of applications in active filters, parasitic cancellation, oscillators, radio frequency (RF) systems etc. [2]. The active simulation of inductors where active devices are employed instead of spiral inductors to emulated the behavior of passive inductors has become popular [1-3]. Analog filters are an integral part of almost every electronic system and so their synthesis and development remains an ever evolving field. Among various filter structures univer- sal filters (UFs) are the most versatile as all the standard filter functions can be derived from them[6-9]. They serve as standalone solution to many filtering needs. The UFs find applications in phase locked loop FM ste- reo demodulator, telephone decoder, speech process- ing etc.[7-8]. The current mode active elements are the best choice for inductor and filter synthesis, own- ing to their large dynamic range, great linearity, wide bandwidth, higher slew rate and excellent low voltage performance compared to their voltage mode coun- terparts [6-7]. Numerous current mode active blocks have been employed for inductance simulation and filter design like second generation current conveyor (CCII) [4], differential voltage current conveyor (DVC- CII) [2], dual X current conveyor (DXCCII) [22], current conveyor transconductance amplifier (CCTA) [28], dif- ferential current conveyor (DCCII) [11], differential dif- ference current conveyor (DDCC) [1], fully differential current conveyor (FDCCII) [34], voltage differencing current conveyor (VDCC) [25], current feedback opera- tional amplifier (CFOA) [13], modified current feedback operational amplifier (MCFOA) [18], inverting current feedback operational amplifier (CFOA-) [5], current dif- ferencing transconductance amplifier (CDTA) [37] etc. The GIS can be categorized based on many criteria like number of active devices used, passive component count, lossy/lossless type inductance realization, use of grounded passive elements, feature of inbuilt tu- nability, capability of active element realization using the off the shelf integrated circuits (ICs) like AD844 etc. The GIS presented in [4, 12-13] require more than one active element for inductance simulation. The GIS imp- lementations presented in [4, 12-13, 15, 21, 23] require more than three passive elements. Although in current fabrication technology it is feasible to implement floa- ting capacitor but use of grounded capacitor has inhe- rent advantages in terms noise cancelation, chip area and parasitic reduction. The GIS implementations in [5, 10, 12, 14-17, 19, 21-24] makes use of one or more floating capacitors. Tunability is one of the major de- sign requirements in today’s mixed mode systems. The GIS circuits in [1, 4-5, 11-24] did not have inbuilt tuning capability. A thorough comparison of the exemplary in- ductance topologies available in the literature with the proposed lossy/lossless GIS topologies is given in Table 1. The literature survey reveals that the draw backs of the majority of the presented inductor simulators are (i) no provision for on chip tunability (ii) use of more than one active element (iii) use of floating capacitor (iv) excessive use of passive elements (v) passive com- ponents matching requirnment. In this research a new active block the extra X current conveyor transconductance amplifier (EXCCTA) is utili- zed in designing six topologies of lossy/lossless indu- ctance simulators. All the designed GIS employ only a single active element and two/three grounded passive elements for implementation. No matching is required for implementation and inductance can be easily tuned via bias current of the transconductor. Further more, two minimum component VM and CM mode universal filters are proposed. The filters structures are characte- rized by following attributes (i) use of minimum num- ber of active blocks and passive components (ii) inbuilt tunability (iii) use of grounded passive elements (iv) provision for independent control of pole frequency and quality factor (v) avalibility of output at low impe- dance in case of VM filter and high impedance in CM filter. The simulation are performed using 0.18μm pa- rameters from TSMC in Spice to validate the theoretical findings. Additionally, an experimental study of pure inductor and VM SIMO filter is conducted using AD844 and LM13700 ICs. The measured results are compared with the ideal ones. 2 Extra X current conveyor transconductance amplifier (EXCCTA) The proposed Extra X current conveyor transconduc- tance amplifier (EXCCTA) is functionally a connection of extra x current conveyor (EXCCII) [26] and operational transconductance amplifier (OTA) [30]. The new block carries features of CCII and tunable OTA in a single in- tegrated circuit structure. The Voltage current charac- teristics of the developed EXCCTA are given in matrix Equation1 and the block diagram is presented in Fig. 1. The block is found to be suitable for implementing minimum component filters and inductor simulators as it provides two independent current input X terminals and an inherent tunable structure. 99 (1) Figure 1: Block diagram of EXCCTA The CMOS implementation of EXCCTA is presented in Fig. 2. It is a 9 terminal active element. The first stage consists of EXCCII transistors (M1-M28). The voltage ap- plied at Y node appears at XP and XN nodes. The cur- rent input at X p node is transferred to nodes Z p+ and Z p- . In the same way the input current from X N node is trans- ferred to Z N+ and Z N- . The current following in Z N and Z p terminals are independent of each other. The class AB output stage is selected for the implementation as it is suitable for low voltage operation [6]. The second stage is composed of OTA. The transconductance is realized using transistors (M29-M40).The output current of the trans-conductor depends on the voltage Z p+ . Assuming saturation region operation for all transistors and equal W/L ratio for transistors M29 and M30 the output cur- rent I o of the OTA is given by Equation 2. () ( ) () 2 Om iZ PB iasi ZP Ig VI KV ++ == (2) Where, the transconductance parameter K i = µC 0x W/2L, (i=29, 30) W is the effective channel width, L is the ef- fective length of the channel, C ox is the gate oxide ca- pacitance per unit area and µ is the carrier mobility. It is evident from (2) that the transconductance can be tuned by the bias current thus imparting tunability to the structure. The most critical parameters of the EXCCTA can be computed by following the analysis procedure given in Table 1: Comparison of exemplary active inductors with the proposed ones Reference No. of Active Devices Used No. of Passive Elements Capacitor Floating Matching Condition Inbuilt Tunability [1] Fig. 5 MDO-DDCC (1) 3 No Yes No [4] CCII (3) 4 No No No [5] CFOA− (1) 3 Yes No No [10] VDBA (1) 2 Yes No Yes [11] DCCII (1) 3 No Yes No [12] OTRA (2) 5 Yes Yes No [13] CFOA (2) 4 No Yes No [14] DCCII (1) 3 Yes No No [15] OTRA (1) 4 Yes No No [16] MICCII (1) 3 Yes Yes No [17] GVCCIII (1) 3 Yes Yes No [18] MCFOA (1) 3 No No No [19] CFOA (1) 3 Yes No No [20] CCII (1)+Current Follower (1) 3 No No No [21] PFTN (1) 5 Yes Yes No [22] DXCCII (1) 3 Yes Yes No [23] OTRA (1)+Voltage Buffer (1) 4 Yes Yes No [24] CFOA (1) 3 Yes No No Fig. 3 Proposed GIS-(1-2, 4-6) EXCCTA (1) 3 No No Yes Fig. 3 Proposed GIS-3 EXCCTA (1) 2 No No Yes M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 100 [6]. The voltage transfer ratio between Y and X p node can be derived as given in Equation 3. () () 0 09 014 91 44 09 014 0 09 014 91 43 09 014 2 1 2 p mm m XP P p Y mm m r rr gg g rr V r rr V gg g rr α + + == ++ + () () 0 09 014 91 44 09 014 0 09 014 91 43 09 014 2 1 2 p mm m XP P p Y mm m r rr gg g rr V r rr V gg g rr α + + == ++ + 4 3 m m g g (3) Where r 0p = r 03 //r 07 , r 0 is the output resistance of the MOS transistor and g m is the transconductance of the MOS transistor. In the similar way the voltage transfer between Y and X N can be obtained as given below. The current transfer ratios β p and β N depend on the load connected at X and Z terminals. This is a slight draw back but if the value of the load connected is less than MOS transistor resistance than an accurate transfer is achieved. The current transfer ratios are derived as giv- en in Equation 5-6. 1 2 m XP N Ym g V Vg α = (4) (5) ZN N XN I I β = 20 24 19 28 mm mm gg gg + + (6) The X terminal resistance of X N and X p terminal is calcu- lated as given in Equations 7-8. () 0 09 014 91 44 09 014 1 2 XP p mm m R r rr gg g rr = ++ + () 0491 4 2 pm mm rg gg + (7) XN R () 0119 28 2 nm mm rg gg + (8) Where r 0p = r 03 //r 07 and Where r 0n = r 02 //r 06 The Z and O nodes impedances are found to be high given by parallel output resistance of MOS transistors. 0100 15 0100 15 ZP rr R rr = + (9) 0200 24 0200 24 ZN rr R rr = + (10) 033 039 0 033 039 Z rr R rr + = + (11) The extra transistor count should not be considered a disadvantage since the unused Z terminal can be re- moved reducing the transistor count. M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 Figure 2: CMOS Implementation of EXCCTA 101 GIS-1 (a) GIS-2 (b) GIS-3 (c) GIS-4 (d) GIS-5 (e) GIS-6 (f) Figure 3: The proposed lossless and Lossy inductor simulators Table 2: The impedance relations of the proposed active inductors Simulator Impedance Realized Inductance ( eq L ) Equivalent Resistance ( eq R ) Type Tunability Matching Requirement GIS-1 eq L 112 m1 2 SCRR g( R+ R) - Pure Yes No GIS-2 eq L 112 m1 2 SCRR g( R+ R) - Pure Yes No GIS-3 eq L 11 m SCR 2g - Pure Yes No GIS-4 11 + eq eq LR 12 m SCR g 1 R L in parallel with R Yes No GIS-5 11 −− eq eq LR 12 m SCR g − 1 R − -L in parallel with -R Yes No GIS-6 + eq eq LR 12 m SCR g 2 1m R Rg L in series with R Yes No M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 102 3 The proposed inductor simulators and universal filters The proposed inductor simulators using a single EXC- CTA and grounded passive elements are shown in Fig. 3(a-f). A simple analysis of the circuits reveals that the GIS-1 to GIS-3 realize pure inductance. The GIS-4 and GIS-5 implements lossy parallel RL inductors. It is note- worthy that the value of the parallel resistor in GIS-4/5 can be set using resistor R 1 independent of the induct- ance value that is set by R 2 and g m which is an advan- tage. Finally, the GIS-6 realizes lossy series RL inductor. The series resistance in GIS-6 can be altered by chang- ing R 1 without affecting the inductance. All the presented inductor simulators use only ground- ed passive elements which is advantageous for inte- grated circuit implementation. The use of grounded capacitor saves chip area and also minimizes noise effects. Furthermore, all the simulators are free from components matching constraints and tunable via bias current of the OTA. The impedance relations of the in- ductors are given in Table 2. Next, the proposed VM universal filter is presented in Fig. 4. The filter structure is obtained by modifying the proposed series inductor GIS-6. The SIMO filter uses ca- nonical number of passive components and can realize low pass (LP), high pass (HP), band pass (BP), notch pass (NP) and all pass (AP) responses. Among the two capaci- tors one is always grounded which is an advantage. The features of the filter includes use of single active block, low output impedance, tunability and provision for in- dependent control of quality factor and pole frequency. The transfer function, expression of -3dB cutoff frequen- cy, quality factor and bandwidth of the filter are given in Equations 12-14. The output can be tapped from Y ter- minal or low impedance X N terminal. The combination of filter input modes and applied input voltages for real- izing all the five filter responses is summarized in Table 3. 2 1222 2 12 3 1 2 1222 2 1 1 mm out mm SCCR SC R VV V gg R V SCCR CR S gg R ++ = ++ (12) 122 m o g CCR ω = (13) 1 1 22 m Cg QR CR = (14) Table 3: The activation sequence of the filter Response Inputs V 1 V 2 V 3 LP 0 0 1 HP 1 0 0 BP 0 1 0 NP 1 0 1 AP 1 -1 1 The MISO CM filter is shown in Fig. 5. The filter employs three grounded passive elements and can realize all LP , BP and HP responses simultaneously. The AP response can be obtained by summing the HP, BP and LP re- sponses. All the responses are available at high imped- ance nodes except the HP response which is available through the capacitor . The HP current can be sensed using a simple current follower. The features of the filter includes use of single active element, grounded pas- sive elements and tunability. The transfer functions for five filter responses are given in Equations 15(a)-15(e). Figure 5: The proposed CM SIMO filter 2 121 21 1 1 LP in m I SCCR I SC R g − = ++ (15a) M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 Figure 4: The proposed VM mode MISO filter 103 2 121 2 121 21 1 m HP in m SCCR g I SCCR I SC R g − = ++ (15b) 21 2 121 21 1 BP in m IS CR SCCR I SC R g = ++ (15c) 2 121 2 121 21 1 1 NP m in m SCCR Ig SCCR I SC R g −− = ++ (15d) 2 121 21 2 121 21 1 1 m AP in m SCCR SC R g I SCCR I SC R g −− + = ++ (15e) 121 1 2 m o g f CCR π = (16) 1 21 m C Q gCR = (17) 4 Non-Ideal analysis In this section the Non-idealities of the EXCCTA are con- sidered and their influence on the proposed inductors and filter circuits is analyzed. A simplified non-ideal model of EXCCTA is presented in Fig. 6 for analysis. The most important aspects contributing to the deviations in frequency performance are the non-ideal frequency dependent current and voltage transfer gains, α i (s) and β i (s) respectively, where α i (s) = α 0i /(1 + s/ω αi ) and β i (s) = β 0i /(1 + s/ω βi ). Ideally, α 0i = β 0i = 1 and ω αi = ω βi = ∞. An- other important performance parameter is the associ- ated parasitics at the X nodes which can be quantified as Z XP = Z XN = R X(N,P) + sL X(N,P) . The parasitic resistance and capacitance associated with the Y and Z nodes are R ZP , R ZN and R Y . While the associated capacitances are C ZP , C ZN and C Y . There ideal values being equal to zero.The γ represents the transconductance transfer inaccuracy of the OTA, while R o and C o are parasitics at the OTA out- put. If the effect of the non-ideal gains is considered the V-I relations of the EXCCTA will be modified to I Y = 0, V XP = β p (s)V Y , V XN = β N (s)V Y , I ZP+ = α p (s)I XP , I ZP- = α p ´(s)I XP , I ZN+ = α N (s)I XN , I ZN- = α N ´(s)I XN , I 0+ = γg m V ZP+ , I 0- = γ´g m V ZP+ . The modified expressions of inductance L eq of the ac- tive inductor simulators taking into account the non- ideal current and voltage transfer gains of the EXCCTA is given in Table 4. Table 4: Modified inductance L eq of the active inductor simulators for non-ideal case GIS-1 () 112 12 eq mP PN N SCRR L gR R γα βα β + ′ = GIS-2 () 112 12 eq mP PN N SCRR L gR R γα βα β + ′ = GIS-3 () 11 eq mP PN N SCR L g γα βα β + ′ = GIS-4 ' 12 1 mPPN N eq g L SCRR γα βα β + ′ = GIS-5 12 1 mPPN N eq g L SCRR γα βα β − =− GIS-6 2 12 mm 1 SCR gg NN eq PP R L R αβ αβγγ =+ ′′ The expressions of transfer function, pole frequency and quality factor including the non-ideal gains for the VM and CM filters are presented in Equations 18-23, re- spectively. 2 22 122 12 3 1 2 22 122 1 N P mm out NN PP mm SC R SCCR VV V gg R V CR SCCR S gg R α αγ γγ αβ αβ γγ ++ = ++ (18) 122 1 2 mPP o g f CCR αβ π = (19) 1 1 22 PP m NN gC QR CR αβγ αβ = (20) 2 ' 121 21 NN LP in NN m I SCCR I SC R g αβ αβ γ − = ++ (21a) 2 121 2 ' 121 21 m HP in NN m SCCR g I SCCR I SC R g γ αβ γ − = ++ (21b) M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 104 21 2 ' 121 21 BP in NN m IS CR SCCR I SC R g γ γ αβ γ ++ ′ = (21c) ' 121 1 2 mNN o g f CCR γα β π = (22) ' 1 21 NN m C Q gC R αβ γ = (23) The sensitivity analysis of both the VM and CM universal filters is also carried out. The Equations 24-26 give the active and passive sensitivities of the VM filter and Equa- tions 27-28 present the sensitivities of the CM filter. 122 1 2 PP m gC CR SSSS SSS ωωωω ωωω αβγ ==== −= −= −= (24) 1 1 NN QQ Q R SSS βα === (25) 12 2 1 2 PP m QQQQQQQ Cg CR SSSSSSS αβ γ =====− =− = (26) ' 121 1 2 Nm N gC CR SSSS SSS ωωωω ωωω βγ α ==== −= −= −= (27) ' 12 1 1 2 Nm N QQQQQQQ Cg CR SSSSSSS βγ α ===− =− =− =− = (28) It can be seen that the active and passive sensitivities are not greater than one which is desirable. Figure 6: Non-ideal model of EXCCTA 5 Simulation results In order to establish the workability of the proposed extra X current conveyor transconductance amplifier (EXCCTA). It was designed in 0.18μm parameters from TSMC. The circuit was simulated in spice to measure the important design metrics. The aspect ratios of the transistors are given in Table 5. The supply voltages are kept at V DD = -V SS = 0.9V. The bias current was fixed at I bias = 100μA. The bias current of OTA was set at 60μA which resulted in a trans- conductance of g m = 0.334mS. The performance param- eters of the EXCCTA are summarised in Table 6. Table 5: Aspect ratios of the transistors Transistor Width (W μm) Length(L μm) M1- M4 3.06 0.72 M5- M7 9 0.72 M9- M23 14.4 0.72 M24- M28 0.72 0.72 M29-M32 3.6 1.8 M33-M40 7.2 1.8 The operation of the inductor simulators is evaluated next. The GIS-1 inductor simulator is tested, the passive component values are selected as R 1 = R 2 = 4kΩ, C 1 = M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 Figure 8: Monte Carlo result of GIS-1for 10% deviation in capacitance value Figure 7: Magnitude and phase response of GIS-1 105 50pF and I bias = 60μA resulting in the inductance of L eg = 0.3mH. The magnitude response of the ideal and simu- lated inductor is given in Fig. 7. To study the effect of pro- cess variability on the performance of the inductor simu- lator Monte Carlo analysis is performed for 10% variation in the capacitance value with Gaussian distribution for 100 runs. It can be inferred from Fig. 8 that there is only slight deviation in the inductance value. To further as- certain the inductor’s performance transient analysis is performed to verify the phase difference between cur- rent and voltage waveforms. The phase difference was found to be 87.2 o as given in Fig. 9 which further verifies the behaviour of the synthetic inductor. Figure 9: Transient analysis of GIS-1 The parallel inductor is now designed. The value of the passive elements are selected as R 1 = R 1 = 4kΩ, C 1 = 50pF and I bias = 60μA resulting in the inductance of L eq = 0.3mH in parallel with R eq = 4kΩ. It is to be noted that by ap- propriately selecting the value of R 1 the value of parallel resistor can be set without affecting the inductance. The magnitude response of GIS-3 is given in Fig. 10. Figure 10: Magnitude response of GIS-3 6 Applications of the proposed inductance simulators To examine the applicability and feasibility of the pro- posed inductor simulators in practical applications they are utilized in numerous applications. The pure inductor simulator GIS-1 is utilized in the design of third order But- terworth ladder filter [5] as shown in Fig. 11. The transfer function of the filter is given in Equation 29. The passive elements of the filter are selected as C 1 = C 2 = 0.1 nF, R s = 1kΩ, R L = 1kΩ and L 1 = 60 uH. The resistors R s and R L are the source and load resistances, respectively. The induc- tor in the passive implementation is replaced by the ac- tive GIS-1 Fig 3(a). The components of GIS-1 are selected as R 1 = R 2 = 4kΩ and C 1 = 10 pF to get L eq = 60uH for the active inductor. The Ideal and simulated filter response are given in Fig.12. It can be deduced that the ideal and simulated curves bear close resemblance. A current mode multifunction filter [27] is also de- signed using the GIS-1. The parallel passive RLC filter is shown in Fig. 13. The filter is designed for -3dB cutoff frequency of 2.054 MHz by selecting components val- ues as C f = 10pF, R f = 10kΩ and L eq = 0.6 mH. The pas- Figure 11: The third order Butterworth filter (29) M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 Table 6: Performance metrics of EXCCTA in Fig. 6 Voltage Gain 1.005 Current Gain ( / ZP XP II + ), ( / ZN XN II + ), 0.9998 Current Gain( / ZP XP II − ), ( / ZN XN II − ), 0.964 Voltage Transfer Bandwidth 756.83MHz Current Transfer Bandwidth ( , PN ZZ ++ ) 774.46 MHz Current Transfer Bandwidth ( , PN ZZ −− ) 711.21 MHz DC Voltage Range ±650mV DC Current Range ( , PN ZZ ) ±140μA DC Current Range ( , PN ZZ −− ) ±60μA X Node Impedance (RX) 39.940Ω X Node Inductance (LX) 2.285μH ( , PN ZZ ) Node Impedance 186.547kΩ ( , PN ZZ −− ) Node Impedance 176.554kΩ 106 sive inductor in the filter is replaced by active inductor GIS-1. The components of GIS-1 Fig. 3(a) are selected as R 1 = R 2 = 4kΩ and C 1 = 100 pF to get L eq = 0.6 mH. The ideal and simulated frequency response of the filter is given in Fig. 14. Figure 14: Frequency response of current mode multi function filter 1 2 o eq f f LC π = (30) The two passive elements based inductor simulator GIS-3 is utilized in the design of voltage mode second order high pass filter as presented in Fig. 15. The trans- fer function and expression for frequency of the filter are given in Equations 31-32. The passive element val- ues for the passive filter are selected as C f = 80 pF, R f = 2kΩ and L eq = 0.6 mH which corresponds to -3dB cutoff frequency of 726.439kHz. To realize the required in- ductance the passive elements of the GIS-3 Fig. 3(c) are chosen to be R 1 = 4kΩ and C 1 = 100 pF to get L=0.6mH for GIS-3. The ideal and simulated response of the filter are given in Fig. 16. Figure 15: Voltage mode HP filter 2 2 11 out in ff eq f V S V SS CR LC = ++ (31) 1 2 o eq f f LC π = (32) Figure 16: Frequency response of Voltage mode filter Now the lossy parallel RL inductor GIS-4 is used in the design of second order current mode HP filter. The pas- sive prototype of the filter is presented in Fig. 17 and the transfer function and pole frequency is given in Equations 33-34. The filter is designed for -3dB cutoff frequency of 918.88 kHz by choosing C f = 50 pF, R eq = 4kΩ and L eq = 0.6 mH. The parallel R eq ∥L eq in the passive prototype are replaced by parallel RL active inductor simulator GIS-4 Fig.3 (d). The GIS-4 is designed with R 1 = R 2 = 4kΩ and C 1 = 100 pF. The ideal and simulated values are given in Fig. 18. To test the signal processing M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 Figure 12: The response of third order Butterworth filter Figure 13: Current mode multi function filter 107 capability of the filter transient analysis is performed. A sinusoidal current signal with 80μA p-p is applied to both filters with ideal passive elements and filter with simulated RL. The results are shown in Fig. 19. It is clear that the ideal and simulated curves follow each other closely verifying the correct functioning of the GIS-4 simulator. 2 2 11 out in ff eq f I S I SS CR LC = ++ (33) 1 2 o eq f f LC π = (34) Figure 17: Current mode high pass filter Figure 18: Frequency response of Current mode high pass filter Figure 19: Transient analysis of Current mode high pass filter 7 Verification of the Proposed Universal Filters The proposed MISO VM filter is tested. The filter is de- signed for -3dB cutoff frequency of 2.054MHz by select- ing C 1 = 20 pF, C 2 = 20 pF, R 1 = 5kΩ and R 2 = 5kΩ. The resulting quality factor of the filter is found to be 1.29. The response of the filter is presented in Fig. 20. The re- sponse of the AP filter is given in Fig. 21. The simulated pole frequency of the is found to be 1.998MHz which translates in to 3.21% error this is due to parasitic ele- ments discussed above. To investigate the signal pro- cessing capabilities of the filter transient analysis is also performed for the BP configuration. It can be inferred from the Fig. 22 that the filter performs well. The total harmonic distortion(THD) of the filter for band pass configuration is also calculated which is found to be perfectly within acceptable limit for wide input voltage range as shown in Fig. 23. The THD remains within 1% till 200mV input voltage. Figure 20: Simulated characteristics of the MISO VM filter Figure 21: Simulated characteristics of the MISO VM AP filter M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 108 The effect of process variations on the proposed filter is studied by performing Monte Carlo analysis of the HP response of the filter. The analysis is carried out for 10% deviation in the two capacitor values adopting the Gaussian distribution. The result for 50 runs given in Fig. 24 shows that the filter functions well with minute deviations that can be accommodated by adjusting the bias current of the OTA for precise frequency control. Figure 24: Monte Carlo analysis of the HP filter for 10% deviation in capacitance values Table 7: Comparison of MISO filters available in the literature with the proposed filter S. No. Type of Filter Active Block Used(No.) No. of R+C No. of grounded C+R Matching Condition Electronic Tunability Inverting Input Needed Low output Impedance [29] MISO (VM) CCII+ (3) 2+2 0+0 No No No (Except for AP) No [30] MISO (VM) DVCC (3) 4+2 2+3 No (Except for AP) No No No [31] MISO (VM) CCTA (1) 2+2 0+0 No Yes No No [32] MIMO (VM) DVCC (1) 2+2 0+0 Yes No No No [33] MISO (VM) DOCCCII (2) 0+2 0+0 No Yes No (Except for AP) No [34] MISO (VM) VD-DIBA 1+2 0+0 No Yes No No [35] MISO (VM) CDBA (2) 4+2 0+0 Yes No No Yes [36] MISO (VM) DDCC (2) 2+2 2+1 No No No No [37] MIMO (VM) FDCCII (1) 3+2 1+1 No No No (Except for AP) No [38] MISO (VM) DDCC (3) 2+2 2+2 No No No Yes Proposed MISO (VM) EXCCTA (1) 2+2 1+0 No Yes No (Except for AP) Yes M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 Figure 23: Total harmonic distortion of the band pass filter configuration Figure 22: Transient analysis of the band pass filter 109 Tunability is an importance feature of filters [28], this at- tribute enables a filter to be useful at different frequencies without requiring any alteration in the passive compo- nents values. The pole frequency of the proposed filter can be adjusted by varying the bias current of the OTA. The LP response of the filter is plotted for different bias currents as shown in Fig. 25. It can be seen from the figure that the bias current slightly effects the quality factor of the filter as can be deduced from Equations (13-14) . To realize a unity quality factor the resistance R 1 should be adjusted accord- Figure 26: Simulated characteristics of the MISO VM BP filter for different quality factors Table 8: Comparison of CM SIMO filters available in the literature with the proposed filter S. No. Type of Filter Active Block Used(No.) No. of R+C No. of grounded C+R Matching Condition Electronic Tunability High output Impedance [39] Fig. 3 SIMO (CM) CBTA (1) 2+2 2+2 No Yes No [40] SIMO (CM) CDTA (1) 2+2 2+0 No Yes No [41] SIMO (CM) DV-DXCCI (1) 2+2 2+2 No No Yes [42] SIMO (CM) UCC (3) 2+2 2+2 No No Yes [43] SIMO (CM) CCCII (3) 0+2 2+0 No Yes Yes Proposed SIMO (CM) EXCCTA (1) 1+2 1+2 No Yes Yes M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 Figure 27: Simulated characteristics of the SIMO CM filter Figure 28: Simulated characteristics of the SIMO CM AP filter Figure 25: Simulated characteristics of the MISO VM LP filter for different bias currents 110 ingly. The quality factor of the proposed filter can also be tuned independent of the pole frequency by varying the resistance R 1 as can be seen in Figure 26 which gives the plot of Q for different values of resistance R 1 . A comparison is provided in Table 7 to compare the proposed VM MISO filter with the other state of the art filter structures. Lastly, the CM SIMO filter is validate by designing it for -3dB cutoff frequency of 2.204MHz by selecting C 1 = 20 pF, C 2 = 20 pF, R 1 = 3.61 kΩ and OTA bias current of 50μA. The frequency response of the filter is given in Fig. 27 and the AP response is shown in Fig. 28 The filter is compared with other exemplary SIMO filter topolo- gies to highlight its advantages as shown in Table 8. 8 Experimental Results The EXCCTA can be easily implemented using commer- cially available integrated circuits AD844 and LM13700. The possible implementation of GIS-1 using only two AD844 and one LM13700 IC is shown in Fig. 29. The ICs are supplied with a bias voltage of ±10V, the I bias of the OTA is fixed at 372µA. The passive components values are chosen to be R 1 = R 2 =1kΩ and C 1 = 10 nF result- ing in the inductance of L=672μH. To test the circuit a triangular waveform with 2V p-p at 17kHz is applied through R 2 and R 3 to convert it in to corresponding in- put triangular current. A square wave form is obtained across the inductor. The oscilloscope output is shown in Fig. 30 which validates the correct functioning of the active inductor. Figure 29: Implementation of the GIS-1 from the com- mercially available ICs To further validate the VM filter it is implemented on breadboard using AD844 and LM13700 ICs. The filter implementation is given in Fig. 31 (a-b). The filter is de- signed for -3dB frequency of 343.199kHz by selecting M. Faseehuddin et al; Informacije Midem, Vol. 48, No. 2(2018), 97 – 113 Figure 31: Experimental setup of the VM filter from the commercially available ICs (a) Block Diagram (b) Bread board implementation (b) (a) passive components values equal to R 1 = R 2 =1kΩ, C 1 = C 2 =1nF and I bias = 232.5µA. The ideal and experimen- tally calculated response of the filter is presented in Fig. 32. The measured frequency of the filter is found to be 321kHz leading to an error of 6.46%, this discrepancy arises due to non-idealities present in the EXCCTA and also due to the parasitics introduced by connecting leads and breadboard, nonetheless it verifies the prac- tical feasibility of the proposed VM filter. Figure 30: The experimental result of the inductor simulator 111 Figure 32: The ideal and experimental results of the VM MISO filter Figure 33: Time domain analysis of LP response (a) 1kHz (b)321 kHz (c) 600 kHz (a) (b) (c) signal of 60mV p-p at frequencies of 1kHz, 321kHz and 600kHz. The filter output obtained on the oscilloscope for LP configuration is presented in Fig. 33 (a-c). 9 Conclusion In this research, a new active element namely EXCCTA is proposed. The EXCCTA is used to design six topolo- gies of lossy and lossless active inductors. Additionally, to further elaborate its versatility two universal VM and CM filter structures employing a single EXCCTA are also proposed. The inductor simulators make use of single EXCCTA, grounded passive elements, requires no pas- sive components matching and are perfectly tunable. The voltage mode filter is obtained by slightly modify- ing the proposed series inductor. The VM filter makes use of canonical number of passive elements and has low output impedance. The CM filter is SIMO type and utilize only three grounded passive elements for imple- mentation. Both the proposed filter structures enjoy low sensitivities, inbuilt tunability and orthogonal con- trol of quality factor and pole frequency. The non-ideal analysis of the inductor simulators and filter structures is also performed. The simulations are performed in 0.18μm parameters from TSMC to verify the theoreti- cal analysis. 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