DESIGN AND DEVELOPMENT OF COMPACT MICROSTRIP ANTENNAS FOR PORTABLE DEVICE APPLICATIONS 1Ahmed Toaha Mobashsher, 2Mohammad Tariqul Islam, 1Norbahiah Misran 1Dept. of Electrical, Electronic and Systems Engineering, Universiti Kebangsaan Malaysia, Bangi, Selangor, Malaysia 2Institute of Space Science (ANGKASA), Universiti Kebangsaan Malaysia, Bangi, Selangor, Malaysia Key words: HCASA, SSA, COMPACT ANTENNA, portable device applications Abstract: During the last decade, wireless communication system developments have been a major motivator of compact antenna research, most particularly for portable device applications. This paper addresses to a development procedure of designing compact antennas. By a careful observation at the current distribution and folding of slots after calculating the resonating slot length, the compactness of microstrip antenna can be achieved. Two compact slot antennas, namely hollow central annular slot antenna (HCASA) and spiral slot antenna (SSA) are designed and prototyped for validation, where the experimental result agrees well the simulation. The overall volume of these antennas are respectively 0.11 1 Xox0.005Xo and 0.08Xox0.08Xox0.005Xo where is the free space operating wavelength, that concludes the reduction of 75 and 86.2% of antenna volume comparing that of an ordinary annular slot antenna operating at the same frequency. Načrtovanje in razvoj kompaktnih mikrostrip anten za uporabo v prenosnih napravah Kjučne besede: HCASA, SSA, kompaktna antena, uporaba za prenosne naprave Izvleček: Razvoj brezžičnih komunikacijskih sistemov je v zadnjem desetletju bil glavna gonilna sila razvoja kompaktnih anten večinoma za uporabo v prenosnih naprav. V članku je opisan postopek razvoja načrtovanja kompaktnih anten do katerih pridemo z analizo tokovne porazdelitve ter rezonančne dolžine in položaja rež. Načrtali in probali smo dva tipa kompaktnih anten z režami, kjer se simulacije ujemajo z eksperimentalnimi rezultati: HCASA -antena s sredinskim obročem in SSA - spiralna antena. Prostornini teh anten so 0.11 X„x0.11 Xox0.005Xo in 0.08Xox0.08Xox0.005Xo, kjer predstavlja valovno dolžino v vakumu; to pomeni zmanjšanje za 75% in 86,2 % volumna navadnih anten, ki delujejo na enaki frekvenci. 1. Introduction In modern mobile and wireless communications systems, there is a increasing demand for compact antennas that can be easily integrated into the portable devices. In this regard, microstrip antennas are highly preferred because of their characteristics such as ease of fabrication and integration, compactness and low profile. Basically the compactness of the antenna is a trade-off between the size and the performance of the antenna due to the fact that antenna performance is bound with the fundamental limits based on the size of the antenna. This is especially true in the field of radio communications, where reducing the size of an antenna leads to smaller and light-weight systems, thereby enhancing portability and minimizing electromagnetic interference with other electronic devices. One way for miniaturization is to alter the geometry of the antenna, such that the electrical length of the current path is increased /1/. Nevertheless, the size of microstrip patch antenna can be reduced with a dielectric of the high relative dielectric constant. But the dielectric caused the degradation of gain and radiation efficiency, thus we need to develop methods for miniaturization through the structural change of the patch. There are some methods reported to minimize the antenna resonating at some lower frequencies. It has been shown that resonant patch antennas can be miniaturized using artificial magneto dielectrics /2/. Further compactness of these antennas has also been achieved through loading, using dielectrics, resistors, shorting pins, or meandering microstrip lines /3/. However, such loadings can increase their loss, complication or fabrication cost. In portable devices there is PIFA, IFA and printed monopole or loop antennas are very promising and widely used as compact antennas /4-6/. However, such internal mobile antennas usually excite large surface currents on the system ground plane of the mobile phone, which functions as an effective radiation portion. An isolation distance of about 7mm or larger between the antenna and the nearby conducting elements or electronic components in the mobile phone is usually required to avoid large degradation effects on the performances of the internal antenna, due to the large excited surface currents on the system ground plane, especially in the region near the internal antenna /7,8/.This is a big limitation for the portable devices. In this paper, a design procedure of compact antennas is proposed which is based on calculating the resonating slot length. The process starts from a conventional annular ring slot antenna and ends with two proposed compact antennas. All of the antennas are designed for operation at 920 MHz. On the basis of the current distribution, it has been shown that the antenna slots can also be folded or coiled in spiral to produce lower frequencies in a low profile by maintaining almost the same slot length. However some impedance matching measures has to be taken to optimize the antennas in desired low frequencies. From the view of simplicity and compactness, these miniaturized antennas are also very prognosticating as they do not need big ground planes to provide good radiation patterns. 2. Theoretical evaluation Slot antennas are attractive because they are easy to analyze, design and fabricate. Their radiation pattern can be bidirectional or unidirectional and is possible to have radiation at low elevation angles for an annular slot of the geometry shown in the co-ordinate system of Figure 1 in an infinite and perfectly conducting ground plane, magnetic surface current of the annular distribution can be given by where is the aperture electric field and n is the unit vector normal to the aperture. The general equations for the-far electric field components Eq and E^ can be written as /9/ Eq --- -M 271 a+W 4n R 0 a ik e-j'^ E^ -.cose. 271 a+W J{M(p)cos(^'-^) + 47t R 0 a where k is the free space propagation constant. Let us consider the point of observation P is represented in the co-ordinate by R, Q and f; the inner radius of the annular slot as a; the width of the slot as Wand where Ef and Ep are respectively the slot electric field components in f and Q directions. For a narrow (W <<10) and fixed annular slot the field component equations can be solved as Slot Fig. 1. Annular-slot Geometry kW Eq = -j"E„a—--—cosn<|)-/;(aÄ:sin0) Z K ff/- ß-M 2 R where J^{aksmQ) = - -jCaksinQ) d(aksmQ) " However, for a constant slot width and when 0.0015 ^ w/ 10 < 0.075 and 3.8 < er < 9.8 an approximation can be provided to obtain the guided wavelength of the resonance by using Bessel functions as /10/ -0.01 In 0.9217- 0.277 ln(er) + 0-0322 /- \ w NO.5 w ^+0.453 ^ h 4.6- 3.65 Nevertheless, the dominant TMn1 mode resonant frequency, fn of the slot antenna is determined by nc fn=- iK^i eff R. m / /2 Here, Rm is the mean radius of the annular ring slot, n is the mode number of resonance, e3ff is the relative efficiency of the slot line and c is the speed of light in free space. Usually, the resonant frequencies are mainly determined by the mean radius of the annular slot. The mean radius can be defined by Rm = (R, + Rg)/2 where, R, is the inner radius and Rg is the outer radius of the annular ring slot. 3. Antenna miniaturization In order to explain the miniaturization technique of the microstrip antenna using current distribution trace, a conventional annular ring slot microstrip antenna has been discussed with a center frequency of 920MHz, shown in Figure 2. The antenna is designed on FR4 substrate of er = 4.4 and tanb =0.02. The antenna dimensions are tabulated in Table I. A symmetrical external 50Q microstrip feeding is introduced to provide impedance matching and excite the conventional annular ring slot. As the fundamental TM11 mode is excited the corresponding equation to determine resonance can be found as - " (a) It is noted that this equation is valid as long as a thin substrate is used for the annular ring antenna. Table 1 Design parameters of the conventional annular slot antenna L W Ro Ri SW FW FL FD 70 70 33 30 3 3 16 35 (a) w b) w c) 0.9 0.92 0.94 Frequency (GHz) Fig. 2. The (a) top view, (b) bottom view and (c) return loss of conventional annular slot antenna b) Fig. 3. Current distribution at 920MHz of (a) conventional annular ring slot antenna and (b) annular ring slot antenna with side feeding As presented in Figure 2(b), the antenna shows a -10 dB bandwidth of 18.2 MHz having central frequency of 920 MHz. In order to understand the effect of the antenna feeding position, the feeding is displaced from the middle to FD = 52.5 mm. Providing proper impedance matching by varying the feeding length, the modified antenna achieves resonance at 920MHz. As seen from the current distribution pattern of the antennas in Figure 3, at resonance frequency, the current densely flows at the edges of the annular slots, while in middle portions of the circular disks the currents are null. The current flows toward the microstrip feeding at the outside portion of the slot and at the inside portion of the slot the current traces flows from the feeding. These two traces produce null current at mean slot length of about 2KRm/2~0.5Xg where guided wavelength at resonance frequency, ^g = ^ /^^eff and effective dielectric constant e^ "'"^^ = 2.7; which makes the current paths symmetrical to the microstrip feeding line /11/. Shifting the antenna feeding line in the upper portion of the antenna increases current density in that region. This introduces the initial step of miniaturization, by cutting the modified antenna in the middle and taking the high c) Fig. 4. Current distribution at 920MHz of (a) half annular ring slot antenna (b) half double annular ring slot antenna and (c) half double annular ring slot antenna without central disk current density portion into account, which is pictured in Figure 4(a). Moreover, in order to match the impedance accurately 2.5mm width is shorten from the half circular antenna, which makes the mean circumference of the antenna =(2-7C •31.5)/2-2-2.5 = 0.47A,g where 1g is the wavelength at the slot at the resonance frequency. This resolves that the minimized slot antenna fed by a microstrip transmission line radiates as a magnetic dipole /12/. Nevertheless, it is exhibited that the current density largely flows on the two edges of the dipole like slot and in the central portion of the antenna current strength gradually decreases. So as the next step of miniaturization, shown in Figure 4(b), another half ring slot of 2mm is cut on the ground plane providing 1mm distance between the slots. This half double annular ring slot antenna (HDARSA) also resonates at the desired frequency of 920MHz. However, this also shows null current traces in the central portion of the disk. So for the sake of miniaturization, in the next step, depicted in Figure 4(c), the half circular disk is etched from the ground and it was observed that the current distribution remains almost similar. However, the length of feeding line is extended for better impedance matching in the resonating frequency. In the following sections two miniaturized antennas are proposed and their optimization process is presented and discussed. 4. Proposed geometry description The configurations of the two compact microstrip slot antennas are illustrated in Figure 5. Both of the antennas are designed on a low cost substrate, FR4 of height hs = hc = 1.6mm with relative permittivity er = 4.4 and loss tangent tanb =0.02. The first antenna is a hollow central annular slot antenna (HCASA), which consists of an annular slot with the central portion etched out from the ground of antenna. And second spiral slot antenna (SSA) having an optimized spiral slot in the ground plane. Both of the antennas are fed by 50Q microstrip lines, which give suitability for the antennas to embed with the circuit boards. The antenna design parameters of the microstrip-fed miniaturized antennas are given in Table II. In Table III the comparative volume of the miniaturized antennas are also mentioned. w. w. (a) b) L, w, (c) d) Fig. 5. Structure of proposed hollow central annular slot antenna (HCASA) top and side view (a & b), spiral slot antenna (SSA) top and side view (c & d) Table 2: Design parameters of the miniaturized antennas Hollow Central Annular Slot Antenna (HCASA) Lc Wc Rci Rc2 SWc SSc FWc FLc FDc e ef 35 35 17 14 3 1 3 10 17.5 45 60 Spiral Slot Antenna (SSA) Ls Ws Rs1 Rs2 SWs SSs FWs FLs FDc 26 26 11.7 2 3 1 3 7 13 Table 3: Comparative volume of the miniaturized antennas Hollow Central Annular Slot Antenna (HCASA) Spiral Slot Antenna (SSA) Antenna Volume (mm) 35x35x1.6 26x26x1.6 Antenna Volume Relative to the Free Space Operating Wavelength (Xo) 0.11 x0.11 X0.005 0.08 X0.08 X0.005 5. Analytical study & optimization The goal of the performed analytical parametric studies is to facilitate more elaboration of the design procedures and optimization processes for miniaturized antennas. Various parameters are investigated to examine the effects of the antenna parameters on resonant frequency, return loss as well as the impedance bandwidth of the antenna. This study covers the formulation of the antenna design, influences of varying the slot lengths on resonant frequency and return loss, the implication of choosing the separation angle for the design, the consequence of changing the offset angle for achieving the best optimized minimized antennas. For better convenience of the effect on the performance of the antenna upon changing the parameters, only one parameter is changed at a time, while keeping others unchanged. 5.1. HCASA The mean circumference of the HCASA slot can be illustrated as, C(0) = 2n.r - Qr Applying this equation for the designed antenna we get, for 0 = n/8 and r = 15.5, the length C(0) = 91. The length of the circular slot is approximately 0.46lg, where 1g is the wavelength at the slot at the resonance frequency. This properly corresponds to the miniaturized half circular antennas discussed previously discussed. Nevertheless, the angles for the two ends of the slots are optimized to achieve proper impedance matching for the best performance of the antenna. Figure 6 shows the relationship between the angle of separation, 0 between the two edges of the slot with the resonant frequency and return loss at the respective frequency. It is seen that the resonant point of the antenna increases almost linearly when the angle, 0 is increased gradually. In the same time, the respective return loss does not behave so monotonously. This is because of the impedance matching of the antenna with feeding strip. In spite of the ripples we can derive a merely straight line for the relation of 0 and return loss. However it is also observed (not shown in graph) that the bandwidth of the antenna increases with increase of the angle. The reason for this phenomenon can be attributed to the relatively electrically large aperture of the antenna as the resonance frequency also increases, having the physical dimensions of the overall patch remaining almost the same. In this design, the angle, 0 = 45° is taken as the optimized one for attaining the resonance frequency of 920MHz. 25 45 65 Separation Angle (6) Fig. 6. The effect of different separation angel, 0 on resonant frequency and return loss 10 20 30 40 50 60 70 Offset Angle (Sy) Fig. 7. The relationship between the offset angle, 0f with resonance and return loss The offset angle of the slot edge, 0f from the line of feeding affects the impedance matching and the resonance frequency dramatically, which is verified by the exhibited graphs of Figure 7 with different values of 0f angle providing other shapes and parameters unchanged. Figure 9 shows that the antenna meets the desired resonance frequency 920MHz when 0f = 10o with a low return loss, which implies a defective impedance matching of the antenna. However the best impedance matching is achieved when 0f = 45o but the resonating frequency is over 920MHz. When 0f = 60o, the resonance falls once again at 920MHz with a better impedance matching and this is taken as the optimized parameter. Nevertheless, with the increasing of when 0f, the resonance point still goes to higher frequencies with a gloomy return loss. 5.2. SSA Spiral arm can be represented as /13/, r(j) = a j + ra, (js < j < je) Here, r(j) is the radial distance from the origin to the arbitrary point on the centre line of the spiral, a is the spiral constant, j is the winding angle, j^ is the spiral start angle, je is the spiral end angle, and ra is the radial distance from the origin to the initial point of the spiral line. The mean arc length of the Archimedean spiral C(j5, je) in polar coordinates between j^ and je is: (1) However, from calculus formula, for the simplification of a variable j representing the angle in radians staring from j =0 radians and wrapping counter clockwise around pole until angle j = je radians it can be derived: l + Cp^iÄp = ^ " _ / 1-\ (pVl+(p + ln cp + vl+cp^